1. Field of the Invention
The present invention relates to an infrared remote control circuit, and in particular, to an infrared remote control circuit that can effectively remove noise.
2. Description of the Related Art
Infrared remote control circuits are commonly used to operate remote electronic or electric equipment by means of infrared rays. For example, they are used to switch the channels of a television receiver.
Referencing FIG. 4, a conventional general reception circuit having an infrared remote control circuit is described. This figure is a block diagram showing a conventional general infrared remote control reception circuit.
An infrared remote control system for remote-controlling the switching over the channels of a television receiver is composed of a transmission section (not shown) having an oscillation circuit and an infrared light emitting diode; and a reception section including an infrared remote control reception circuit such as that shown in FIG. 4.
The transmission section oscillates a pulse position modulation (hereafter simply referred to as "PPM") signal provided by discontinuing a carrier of a specific frequency in order to operate the infrared LED to transmit the PPM signal to the reception section as an infrared modulation wave that uses infrared rays as a medium.
The reception section normally comprises an infrared sensing element 1 consisting of a Pin photodiode; an amplifying circuit 2; a band pass filter (hereafter simply referred to as a "BPF") 3 that tunes with a carrier for the PPM signal; a wave detection circuit 4, a waveform shaping circuit 5 including a hysteresis comparator; and an output terminal 6 in order to output a pulse signal depending on whether the carrier for the PPM signal is interrupted.
The PPM signal, which is transmitted as the infrared modulation wave, is received by the infrared sensing element 1 and amplified by the amplifying circuit 2 using an appropriate gain. The BPF 3 removes unwanted signal or noise from the amplified signal, and the wave detection circuit 4 detects a low or a high level depending on whether the carrier for the PPM signal is interrupted. The hysteresis comparator in the waveform shaping circuit 5 shapes an output signal from the wave detection circuit 4, which is then output from the output terminal 6 as a pulse signal that is output depending on whether the carrier for the PPM signal is interrupted.
Referencing FIG. 3, the configuration of a part of the conventional infrared remote control circuit that is located after the BPF 3 is described in detail.
In the BPF 3, a first capacitor C1.multidot.7 is connected at one end to an output terminal of the amplifying circuit 2 and at the other end to an input terminal of a first buffer circuit 12 and an output terminal of a first variable transconductance amplifier 11. The first variable transconductance amplifier 11 has a non-inverted and an inverted input terminals.
The output of the first buffer circuit 12 is connected to a non-inverted input terminal of a second variable transconductance amplifier 13 having a non-inverted and an inverted input terminals. An output terminal of the second variable transconductance amplifier 13 is connected to an input terminal of a second buffer circuit 14 and is grounded via a second capacitor C2.multidot.8.
An output terminal of the second buffer circuit 14 is connected to an input terminal of the wave detection circuit 4 and to the inverted input terminals of the first and second variable transconductance amplifiers 11 and 13. The non-inverted input terminal of the first variable transconductance amplifier 11 is connected to a positive terminal of a voltage source 113. An output terminal of a current mirror circuit 19 is connected to the first and second variable transconductance amplifiers 11 and 13 to allow currents I1 and I2 to flow as control signals.
The above circuit constitutes the BPF 3 having one end of the first capacitor C.multidot.17 as a signal input terminal and the output terminal of the buffer 14 as a signal output terminal.
gm (transconductance) of the first and second variable transconductance amplifiers 11 and 13 used for the BPF 3 is expressed by the following Equation (1).
[Equation 1] ##EQU1##
K=Boltzmann's constant PA1 T=Absolute temperature PA1 q=Amount of charges in electrons PA1 RE=Value of resistors R1 and R2 PA1 I1=Current value of an control signal from the first variable transconductance amplifier 11 PA1 I=Current value of an control signal from the second variable transconductance amplifier 13 PA1 f1: PPM signal carrier frequency PA1 C3: Capacitance value of the capacitor C3 ##EQU4## PA1 f1: PPM signal carrier frequency PA1 C3: Capacitance value of the capacitor C3 PA1 another wave detection circuit is provided between the first wave detection circuit and the waveform shaping circuit as a second wave detection circuit, and in that the output from the same current mirror circuit is used as a control signal for both BPF and second wave detection circuit.
gm decreases with increasing I1 (or decreasing I2) while it increases with decreasing I1 (or increasing I2). Hereafter, a lead-in terminal of I1 is referred to as a negative control terminal and a lead-in terminal of I2 is referred to as a positive control terminal of the variable transconductance amplifier. PA2 gm of the variable transconductance amplifiers 11 and 13 can be varied by setting the voltage from the voltage source 113 to fix I1 to an appropriate value while varying the value of the variable resistor R3 to vary the value of I2.
When the capacity values of the first and second capacitors C1.multidot.7 and C2.multidot.8 are designated as C1 and C2, respectively, and gm of the variable transconductance amplifiers 11 and 13 are indicated as gm1 and gm2, respectively, the tuning frequency f.sub.0 (hereafter referred to as f.sub.0) of the BPF 3 shown in FIG. 3 can be expressed by Equation (2).
[Equation 2] ##EQU2##
The tuning frequency f.sub.0 of the BPF 3 can be adjusted by using the variable resistor R3 to control the lead-in current I2 at the positive control terminal of the second variable transconductance amplifier 13.
Conventional infrared remote control reception circuits are generally composed of semiconductor integrated circuits. During an impurity diffusion step in a semiconductor integrated circuit fabrication process, the diffusion of impurities more or less varies, resulting in differences in the values of resistors and capacitors in the semiconductor integrated circuit constituting the infrared remote control circuit. As a result, the tuning frequency f.sub.0 of the BPF 3 varies.
As the resistance value varies, the value of I1 on the circuit varies. Since, however, I1 contributes as a product with RE, which is the resistance value of the resistors R1 and R2, as shown in Equation (2) defining f.sub.0, it does not substantially vary the value of f.sub.0. On the other hand, the variation of the value of I2 directly noticeably varies the value of f.sub.0.
Thus, the variable resistor R3 that determines I2 is not provided on the semiconductor integrated circuit but outside it, or if it is provided on the semiconductor integrated circuit, trimming is carried out so that f.sub.0 remains unchanged despite the difference in the values of the internal resistors of the semiconductor integrated circuit.
In addition, f.sub.0 directly varies if the capacities of the capacitors C1.multidot.7 and C2.multidot.8 differ. To deal with this problem, the resistor R3 has a variable resistance so that the variation of f.sub.0 is compensated for by adjusting the resistance value of the resistor R3 after the diffusion of impurities.
Next, the configuration of the wave detection circuit 4 is described. The output terminal of the BFP circuit 3 is connected to a base of an NPN transistor Q100 and an input terminal of a DC level shift circuit 15. The output of the DC level shift circuit 15 is connected to an input terminal of a low pass filter 16, and an output terminal of the low pass filter 16 is then connected to a base of an NPN transistor Q101.
Emitters of the NPN transistors Q100 and Q101 are connected to output terminals 23.2 and 23.3 of a current mirror circuit 23, respectively. A collector of the NPN transistor Q100 is connected to Vcc, and a collector of the NPN transistor Q101 is connected to an input terminal 17.1 of a current mirror circuit 17.
An output terminal 17.2 of the current mirror circuit 17 is connected to the input terminal of the waveform shaping circuit 5 and the output terminal 23.3 of the current mirror circuit 23, and is grounded via a third capacitor C3.
The above circuit constitutes the wave detection circuit 4.
Next, referencing FIG. 5, the operation of the wave detection circuit 4 is described. FIG. 5(a) shows an example of a PPM signal waveform composed of a first and a second ON time periods having a carrier and a first and a second OFF time periods having only a DC signal. The pulse in the second OFF time period is not a signal but a noise.
The PPM signal shown in FIG. 5(a), which is output by the BPF 3, is input to the wave detection circuit 4 and branches into two paths. One of the paths directly leads to the base of the NPN transistor Q100, and the other path passes through the DC level shift circuit 15, which applies a DC offset to the signal, and then the low pass filter 16, which removes the carrier from the signal, and finally leads to the base of the NPN transistor Q101. FIG. 5(b) shows the waveforms of signals input to the bases of the NPN transistors Q100 and 101, respectively.
The NPN transistors Q100 and Q101 operate as differential switches. When the base potential of the NPN transistor Q100 is lower than the base potential of the Q101, the NPN transistor Q101 is turned on to cause a current to flow through the output terminal 17.2 of the current mirror circuit 17. On the other hand, when the base potential of the NPN transistor Q100 is higher than the base potential of the Q101, the NPN transistor Q101 is turned off to prevent currents from flowing through the output terminal 17.2 of the current mirror circuit 17.
By appropriately increasing a current I4 (hereafter simply referred to as "I4") that starts to flow from the current mirror circuit 17 when the NPN transistor Q101 is turned on, beyond a current I3 (hereafter simply referred to as "I3") flowing through the output terminal 23.3 of the current mirror circuit 23, the capacitor C3 is charged with the differential current between I4 and I3 when the NPN transistor Q101 is turned on, while it is discharged with I3 when the NPN transistor Q101 is turned off.
While the PPM signal is turned on, the charging current that equals the difference between I4 and I3 is higher than the discharge current from I3, so the capacitor C3 provides a high level while repeating charging and discharging in a sawtooth waveform. While the PPM signal is turned off, the capacitor C3 provides a low level using only the discharge current from I3. The charging and discharging voltages are shown by Equations (3) and (4). ##EQU3##
FIG. 5(c) shows a voltage waveform of the charging and discharging of the third capacitor C3.
A charging and discharging signal from the capacitor C3 is input to the waveform shaping circuit 5, which shapes the waveform by setting the hysteresis width of the hysteresis comparator 18 so that it does not respond to the crest value of the sawtooth waves. Then, the pulse signal shown in FIG. 5(d) which is proportional to the ON time period of the PPM signal is output from the output terminal 6.
In this case, the capacitor C3 is charged and discharged at the voltage defined by Equations (3) and (4). However, when the potential at the capacitor C3 rises to cause the sawtooth wave to exceed the threshold of the hysteresis comparator 18 and if it repeatedly exceeds the threshold, the waveform is broken to cause malfunction.
Thus, conventionally, the same current mirror circuit supplies both the current that defines the carrier frequency f1 (hereafter referred to as "f1") of the PPM signal and the current that defines the hysteresis width, thereby determining the hysteresis width so as to prevent malfunction even if the resistance and capacitance of resistors and capacitors are set at different values during the fabrication of the semiconductor integrated circuit.
The conventional infrared remote control reception circuit has the following problem. Optical noise from an invertor fluorescent lamp or noise in a horizontal-synchronization signal to a television receiver at around 15 kHz appears, as shown in FIG. 5(a), at the output terminal of the BPF 3 as a short noise during the second OFF time period. Then, as shown in FIG. 5(c), this noise appears in the charging and discharging voltage waveform of the capacitor C3 and exceeds the threshold of the hysteresis comparator 18, causing malfunction in which the output is inverted during the OFF time period of the PPM signal as shown in FIG. 5(d).
As a technique for preventing such malfunction, Japanese Patent Application Laid-Open No. 60-141037 and 60-141038 has proposed a circuit such as that shown in FIG. 7.
In the circuit shown in FIG. 7, a noise elimination circuit 17 at which an infrared signal pulse continuously arrives at least twice and which provides output if the pulse interval is longer than or equal to a predetermined value is provided at the output side of the wave detection circuit 10 that detects infrared signal pulses. The noise elimination circuit 17 is provided with a charge/discharge circuit 18 for charging or discharging the capacitor 16 according to the detection output. The output of the charge/discharge circuit 18 is connected to a comparator 20 the output of which is inverted when the terminal voltage of the capacitor 16 exceeds a predetermined level.
According to this circuit configuration, if the carrier frequency f1 input from the terminal 4 continuously arrives as a signal that comprises at least two pulses with a pulse interval longer than or equal to a predetermined time period, a charging and discharging circuit 18 charges and discharges a capacitor 16 in response to a wave detection output. When the voltage at the terminal of the capacitor 16 exceeds a predetermined level, the output from a comparator 20 is inverted.
The time constant for the charging and discharging of the capacitor 16 is set so that discharging is faster than charging if at least two infrared signal pulses of the carrier frequency f1 continuously arrive and if the pulse interval is shorter than or equal to a predetermined time period. This setting enables signals with reduced noise to be output to the output terminal, as shown in FIG. 8D.